Current steering circuit, corresponding device, system and method

ABSTRACT

A circuit includes a first transistor and a second transistor having respective control terminals coupled to receive first and second bias voltages. A first electronic switch is coupled in series with, and between current paths of the first and second transistors to provide an output current line between a circuit output node and ground. A second electronic switch is selectively activated to a conductive state in order to provide a charge transfer current path between a bias node and a charge transfer node in the output current line. A third electronic switch is selectively activated to a conductive state in order to provide a charge transfer current path between the charge transfer node and the control terminal of the second transistor.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.16/221,855 filed Dec. 17, 2018, which claims the priority benefit ofItalian Application for Patent No. 102017000148935, filed on Dec. 22,2017, the contents of which are hereby incorporated by reference intheir entireties to the maximum extent allowable by law.

TECHNICAL FIELD

The description relates to steering current Digital to Analog Converter(DAC) design, for use e.g. in Laser Diode Drivers (LDD).

BACKGROUND

Laser diodes for pico-projector applications are conventionally driventhrough a programmable current source, e.g. a DAC. Rise/fall times andsettling time of an output current are comprised in parameters that maybe used to define the quality of the projection, and designing the DACsin order to meet these specifications is desired.

Conventional DAC circuits may comprise multiple output currentterminals, e.g. a positive and negative output terminal, which may becontrolled by corresponding switching transistors, which may selectivelycouple the output current terminals to a current source, e.g. cascodedcurrent source transistors. The switching transistors may be selectivelyactivated and deactivated, by means of a digital control signal, toprovide current at the positive and negative output terminals.

The switching transistors may selectively couple the positive andnegative output terminals to the current source, e.g. through a commonnode.

It has been observed that such conventional circuits, however, mayexhibit one or more of the following drawbacks:

-   -   the control terminals (e.g. gates) of the current source        transistors may require a constant bias voltage applied thereto,        thereby keeping a constant current flow,    -   if both the switching transistors are simultaneously OFF, the        common node connecting the current paths of the switching        transistors is (rapidly) discharged and it may require a        non-negligible period of time to recharge; accordingly, the        switching transistors are driven in order to avoid that both        switching transistors are simultaneously off,    -   there is a constant DC power dissipation, and/or    -   turn on/off of the switching transistors may cause a glitch in        the output current, due to stray capacitances active between the        control terminal of each switching transistor and the        corresponding output terminal coupled thereto, e.g. between gate        and drain in case of MOSFET switching transistors.

Such a conventional circuit was modified, and further solutions areknown for example from document H. Takakura et al.: “A 10 bit 80 MHzglitchless CMOS D/A converter”, IEEE, 1991, Custom Integrated CircuitsConf. (CICC), incorporated by reference. It was observed that thesolution disclosed by Takakura, comprising switching transistors, aimsto reduce a current feedthrough occurring, due to the straycapacitances, between the control terminals of the switching transistorsand corresponding output terminals coupled to the switching transistors.Also, a reduction of switching noise caused by the switching transistorsmay be achieved and a settling time of the DAC converter may beimproved. However, it was observed that the solution may lead to areduced output voltage swing.

Also, U.S. Pat. No. 7,138,855 (EP 1366568 B1), incorporated byreference, teaches a circuit where, during power-off transients of anoutput current, a stray capacitance may couple a step-down voltage to anode coupled to a current source transistor. In such case, the step-downvoltage increases a voltage regulating the current passing through thecurrent source transistor (e.g. increases the gate-source voltage incase of FET transistors) and produces a spike in the output current,with the generated spike in the current that discharges the straycapacitance.

U.S. Pat. No. 7,138,855 aims at providing an improved solution withrespect to circuits according to the prior art, for switching a currentoff without inducing an overcurrent event and discloses, in order toreduce occurrence of overcurrent, an arrangement wherein a switch isintroduced parallel to the capacitance, which, if active, short circuitsand discharges the stray capacitance. The discharge current, therefore,does no longer produce a current spike in the output current and is nottransmitted to a load.

Also, if the bias voltage applied to the control terminal of the currentsource transistor is lower than the sum of the threshold voltages of thecurrent source transistor and a switching transistor, then the voltageregulating the current passing through the switching transistor (e.g.the gate-source voltage thereof) may reach the threshold voltage, and ifsuch voltage is lower than the threshold voltage of the switchingtransistor then the current source transistor switches off.

Ka-Hou Ao Leong et al., “Design of a 1-V 10-bit 120-MS/sCurrent-Streering DAC with Transient-Improved Technique”, RIUPEEEC,2006, incorporated by reference, discloses a cascode-switch currentsource (CSCS) circuit. The Leong reference aims at reducing a distortionwhich may be due to a falling settling time being different than arising settling time for an output current. It was observed that:

-   -   when the switching transistors present in the document are off,        respective nodes, coupled to their current paths, may be        exponentially charged until further transistors, similarly        coupled to the respective nodes, enter in the subthreshold        region;    -   stray capacitances may be also present at the respective nodes,        which may further extend the transient(s); and    -   a Charge-Removal-Replacement (CRR) topology provides the charge        that may be required to settle faster the respective nodes.

Also, it was observed that one or more timing advantages due to a bettersettling of the node may be present, e.g.:

-   -   a balanced speed and advanced transition: due to the absence of        an exponential discharge for the cascode, a balanced falling and        rising speed may be obtained;    -   recovered synchronization for each current source: the        respective nodes may require same settling time (charging time)        during each clock cycle, thus no delay difference among active        current paths and no input-data-dependent non-linearity are        present; and    -   susceptibility to asynchronous glitches may decrease.

However, it was observed that the circuit may exhibit one or moredisadvantages, e.g. precise CRR capacitances may be desired.

Further DAC circuits were observed, for example a circuit exemplified indocument U.S. Pat. No. 5,548,238, incorporated by reference, which aimsat reducing consumption in idle mode. In conventional DACs, an idlecurrent is steered to ground, thereby leading to a high powerconsumption in idle mode. In the document, however, the current path maybe switched off, thereby possibly achieving no power consumption in idlemode.

In the exemplified DAC, it may be possible to turn on a current sourcetransistor in order to establish a stable current and to bias all thenodes related thereto, before turning to an “on” mode the DAC. In orderto achieve a smooth transition, the current steering circuitry may beturned to the “on” mode after the current source transistor is turnedon.

Conversely, the transition from “on” to “off” mode is achieved by firststeering the current to ground and then cutting off the current sourcetransistor.

It was also observed that two capacitors may be introduced in the DACcircuit, in order to compensate the distortion on the bias voltagescaused by coupling with stray capacitors (e.g. stray capacitors betweendrain and gate of MOSFET transistors, e.g. the current sourcetransistor) during the current source switching.

SUMMARY

There is a need in the art for a current steering DAC that meets one ormore of the following characteristics: high dynamic performance withsharp rise/fall times (in the order of hundreds of picoseconds—1ps=10⁻¹² s); high speed switching rate with a fast settling time (10bits resolution up to 300 MHz sampling frequency); high current swingwith output full scale programmed at 10 bits resolution; and/or lowpower with substantially no power consumption for the non-selectedgenerators of the DAC.

One or more embodiments may comprise a corresponding device (e.g. anLDD—laser diode driver device, e.g. for pico-projector applications), acorresponding system (e.g. laser diode driver devices plus laser diodes)and a corresponding method of driving a circuit according to one or moreembodiments.

IN an embodiment, a circuit comprises: a first transistor and a secondtransistor having respective control terminals and current pathstherethrough, the control terminal of the first transistor coupled to afirst bias voltage node and the control terminal of the secondtransistor coupled to a second bias voltage node; a first electronicswitch having a selectively activatable current path therethrough;wherein the current paths through the first and second transistor andthe current path through the first electronic switch are cascaded in anoutput current line between a circuit output node and ground, the outputcurrent line having an intermediate portion between the first and secondtransistor with a charge transfer node in the intermediate portion.

The circuit may comprise: a second electronic switch arranged between athird bias voltage node and the charge transfer node, and a thirdelectronic switch arranged between the charge transfer node and thecontrol terminal of the second transistor, the second electronic switchand the third electronic switch selectively activatable to a conductivestate to provide charge transfer current paths coupling the chargetransfer node to the third bias voltage input node or the second biasvoltage input node, respectively.

In one or more embodiments, the first electronic switch is arranged inthe portion of the output current line between the first transistor andthe second transistor.

In one or more embodiments, the first electronic switch is arranged in arespective portion of the output current line between the circuit outputnode and the second transistor or between the first transistor andground.

In one or more embodiments, the circuit comprises at least one of: afirst capacitor between the control terminal of the first transistor anda node sourcing an inverted replica of the control signal, and/or asecond capacitor between the control terminal of the third transistorand a node sourcing the control signal.

In one or more embodiments, the first and second transistor and thefirst, second and third electronic switches comprise field effecttransistors, such as MOSFET transistors, having gate control terminalsand source-drain current paths therethrough.

Also, in one or more embodiments, the first and second transistor andthe first, second and third electronic switches comprise field effecttransistors having their bulk terminals coupled to ground.

In an embodiment, a device comprises: at least one circuit as disclosedabove which is configured to provide an output current via the outputcurrent line between the output node and ground of the at least onecircuit, and at least one driver circuit coupled to the at least onecircuit to provide the control signal to selectively activate the firstswitch as well as activation signals to selectively activate the secondswitch and the third switch.

In an embodiment, a system comprises: the device as disclosed above, andat least one user component, optionally a laser diode, coupled to thedevice to receive therefrom the at least one output current.

One or more embodiments may further relate to a method of driving any ofthe circuit, the device or the system as disclosed above. The methodcomprises: selectively activating the first electronic switch to theconductive state synchronously with an on-off control signal havingrising edges and falling edges, and selectively activating the secondelectronic switch and the third electronic switch to the conductivestate synchronously with the rising edges and falling edges of theon-off control signal selectively activating the first electronicswitch.

Also, the method comprises: activating the second electronic switch tothe conductive state at the rising edges of the on-off control signalselectively activating the first electronic switch, and activating thesecond electronic switch to the conductive state at the falling edges ofthe on-off control signal selectively activating the first electronicswitch.

The method further comprises maintaining the second electronic switchand the third electronic switch to a non-conductive state over aninterval between the rising edges and the falling edges of the on-offcontrol signal selectively activating the first electronic switch.

BRIEF DESCRIPTION OF THE DRAWINGS

One or more embodiments will now be described, by way of example only,with reference to the annexed figures, wherein:

FIG. 1 is exemplary of a circuit according to one or more embodiments,

FIG. 2 represents driving and output signals according to one or moreembodiments, and

FIG. 3 is exemplary of a device comprising one or more circuitsaccording to one or more embodiments.

DETAILED DESCRIPTION

In the ensuing description, one or more specific details areillustrated, aimed at providing an in-depth understanding of examples ofembodiments of this description. The embodiments may be obtained withoutone or more of the specific details, or with other methods, components,materials, etc. In other cases, known structures, materials, oroperations are not illustrated or described in detail so that certainaspects of embodiments will not be obscured.

Reference to “an embodiment” or “one embodiment” in the framework of thepresent description is intended to indicate that a particularconfiguration, structure, or characteristic described in relation to theembodiment is comprised in at least one embodiment. Hence, phrases suchas “in an embodiment” or “in one embodiment” that may be present in oneor more points of the present description do not necessarily refer toone and the same embodiment. Moreover, particular conformations,structures, or characteristics may be combined in any adequate way inone or more embodiments.

The references used herein are provided merely for convenience and hencedo not define the extent of protection or the scope of the embodiments.

A solution according to one or more embodiments may improve rise/falltimes and settling time of an output current of a steering currentdigital-to-analog converter, DAC, suitable to operate with a largeoutput full-scale range and without consumption in case of non-selectionof the current generator itself.

One or more embodiments, exemplified in the following figures, mayrelate to high-speed, high-power and high current swing switched-cascodecurrent mirrors.

One or more embodiments may relate to a current steering cell exhibitingone or more of the following features:

-   -   a high output FSR—full scale range—current swing (e.g.        I_(OUT,FSmax) being about 1000*I_(OUT,FSmin)): the output        current IOUT may be programmed accordingly to DAC input data,        e.g. from 0 to a full-scale value with a given, i.e. 10b,        resolution, the full-scale value may be also independently        programmed between a minimum and a maximum value,    -   a high switching rate (FSW—switching frequency of about 300        MHz),    -   low rise/fall time (T_(rise/fall) of about 500 ps), and    -   low power consumption.

FIG. 1 exemplifies a circuit 10 according to one or more embodiments.The left portion of FIG. 1, represents a possible implementation of abias circuit 1 producing first, second and third bias voltages to thecircuit 10 at respective first, second and third bias voltage nodes VB1,VB2 and VB3.

The bias circuit 1 may comprise a reference current line, between acurrent reference node Iref, to which a desired output FSR current maybe applied, and ground GND.

A first, second and third bias transistors B1, B2 and B3 may be present,having control terminals (e.g. gate terminals in case of FETtransistors) coupled to the current reference node Iref, a first voltagenode Vcc and a second voltage node Vb, respectively. The voltage nodesmay apply DC biasing signals to the control terminals of the biastransistors.

Also, current paths through the first, second and third bias transistorsB1, B2 and B3 may be arranged cascoded in the reference current linebetween the current reference node Iref and ground GND.

The first bias voltage node VB1 may be coupled to a buffer stage BS,which in turn may be connected (e.g. directly) to the control terminalof the first bias transistor B1. The second bias voltage node VB2 may becoupled to a buffer stage BS, which in turn may be connected (e.g.directly) to the control terminal of the third bias transistor B3. Thethird bias voltage node VB3 may be coupled to a buffer stage BS, whichin turn may be connected (e.g. directly) to a node B′ of the currentline, arranged intermediate of the current paths of the second and thirdbias transistors B2 and B3.

The bias voltages may be therefore provided to the circuit 10, whereinthe bias circuit 1 may provide a “replica” of the current generatorbranch of circuit 10. For example, the third bias voltage provided atthe third bias voltage node VB3 may provide a replica of the voltage atan intermediate node (e.g. a second intermediate node Y) when thecurrent generator of the circuit 10 is active.

In FIG. 1, an output current line between an output current node Ioutand ground GND may be present, wherein the current path of a currentsource transistor M1, an electronic switch (e.g. a switching transistor)M2 and a cascode transistor M3 may be arranged in the output currentline. Each transistor may have a control terminal plus a current paththerethrough.

In one or more embodiments, the current source transistor M1 may beconfigured for generating an output current, the electronic switch M2may be configured for controlling the current source (for switching iton and off) and the cascode transistor M3 may be configured forimproving an output impedance and, e.g. in case of use with high-voltagedevices, for facilitating achieving protection against a voltage at theoutput current node Iout that may overcome a Safe-Operating-Area for theremaining components in the output current line of the circuit 10, e.g.M1, M2.

For example, for applications in the field of current generators forlaser drivers, the voltage at the output current node Iout may rise to ahigh voltage (e.g. 10-12 V) that may depend on certain features of laserdiodes, for example on forward voltage, power supply and/or inductiveringing thereof. However, to facilitate achieving high switching speeds,the current generator active devices (e.g. transistor M1, switch M2) maybe designed with certain technological options that may lead to suchdevices not being able to sustain high voltages and the cascodetransistor M3 may be used to provide protection with respect to the highvoltage on the output current node Iout.

The current source transistor M1, the electronic switch M2 and thecascode transistor M3 may be arranged cascoded, i.e. with current pathstherethrough coupled in series.

In one or more embodiments, the current path of the current sourcetransistor M1 may be connected to ground GND and to a first intermediatenode X, the current path of the electronic switch (e.g. a switchingtransistor) M2 may be arranged in an intermediate portion of the outputcurrent line between the current source transistor M1 and the cascodetransistor M3, e.g. the switching transistor M2 may be connected to thefirst intermediate node X and to a second intermediate node Y, and thecurrent path of the cascode transistor M3 may be connected to the secondintermediate node Y and the output current node Iout.

Conversely, in one or more embodiments, the current path of theelectronic switch (e.g. a switching transistor) M2 may be arrangedbetween the current source transistor M1 and ground GND, or optionallybetween the output current node Iout and the cascode transistor M3. Insuch embodiments, the first intermediate node X and the secondintermediate node Y may be (e.g. directly) connected.

For example, providing the electronic switch M2 between the currentsource transistor M1 and the cascode transistor M3 may facilitateovercoming matching and speed limitation issues.

It will also be appreciated that, to comply with the specifications thatmay be desired for the output current Iout, e.g. a power consumption,the current source transistor M1 and the cascode transistor M3 (plusoptionally the switching transistor M2) may be sized according to aminimum available headroom voltage and a maximum full-scale. As aconsequence, in case minimum full-scale is selected, the current sourcetransistor M1 and the cascode transistor M3 may operate inweak-inversion, and large parasitic capacitances associated thereto maybe charged and discharged with the low currents. The stray (andparasitic) capacitances may be relevant to operation of the circuit 10and may affect dynamic performances (rise/fall and settling time of theoutput current).

In one or more embodiments, the control terminal of the current sourcetransistor M1 may be coupled to the first bias voltage node VB1, and thecontrol terminal of the cascode transistor M3 may be coupled to thesecond bias voltage node VB2. Also, the electronic switch M2 mayselectively switch to a conductive state as a function of a controlsignal C. For example, in case the electronic switch M2 may comprise aswitching transistor, the control terminal of the switching transistorM2 may be (e.g. directly) connected to a control node, providing thecontrol signal C. The control signal C may include an on-off signal,e.g. a digital signal switching between a high value (e.g. 1) and a lowvalue (e.g. 0).

In one or more embodiments, current paths of a first M4 and second M5active bootstrap switches (in the non-limiting example of FIG. 1represented as FET transistors) may be arranged between the second biasvoltage node VB2 and the third bias voltage node VB3. The current pathsof the active bootstrap switches M4, M5 may be connected (e.g. directly)to a third intermediate node Z, with the node Z short circuited to thesecond intermediate node Y. The first M4 and second M5 active bootstrapswitches may be selectively activatable by a first monostable signal BC1and a second monostable signal BC2, respectively, which may follow theon/off switching of the control signal C.

In one or more embodiments, wherein the first M4 and second M5 activebootstrap switches may comprise switching transistors, a firstmonostable signal BC1 and a second monostable signal BC2 may be appliedto the control terminals thereof, respectively.

In one or more embodiments, a first C1 and second C2 capacitor may beintroduced in order to compensate distortion on the bias voltages, e.g.the first and second bias voltages, that may be caused by straycapacitors Cgd and Cgs (active on the current source transistor M1 andthe cascode transistor M3), respectively, during theactivation/deactivation of the current source. The first capacitor C1and the second capacitor C2 may be used to compensate at first order acharge sharing related to charge/discharge of the stray capacitances Cgdand Cgs. The charge sharing related to the stray capacitors Cgd and Cgsmay interfere with the transient of the output current Iout andoperation of the current source transistor M1 and the cascode transistorM3, respectively.

The current source transistor M1 may have a stray capacitance Cgdbetween its control terminal and its current path, at the firstintermediate node X towards the cascode transistor M3, and the cascodetransistor M3 may have a second stray capacitance Cgs between itscontrol terminal and its current path at the second intermediate node Ytowards the current source transistor M1.

In one or more embodiments, the first capacitor C1 may be arrangedbetween the control terminal of the cascode transistor M3 and to acontrol signal C node, and the second capacitor C2 may be arrangedbetween the control terminal of the current source transistor M1 and toa node reproducing an inverted replica C of the control signal C.

Accordingly, the control signal C may be applied to the electronicswitch M2 (e.g. to the control terminal of the switching transistor M2)plus to the second capacitor C2, while an inverted replica C of thecontrol signal C may be applied to the first capacitor C1. Also, themonostable signals BC1 and BC2, which may be synchronized to the controlsignal C, may be used to selectively active the first and secondbootstrap switches M4 and M5, respectively.

In one or more embodiments, the active bootstrap switches, M4 or M5, maybe selectively activatable to a conductive state to provide chargetransfer current paths coupled to the first and second straycapacitances Cgs, Cgd by coupling the second intermediate node Y to thethird bias voltage node VB3 or the second bias voltage node VB2,respectively.

FIG. 2 is a non-limiting example of a behavior over time of the outputcurrent signal I, the control signal C plus monostable signals BC1 andBC2.

In one or more embodiments, the control signal C and the monostablesignals BC1 and BC2 may be on-off digital signals, e.g. switchingbetween a high value (e.g. 1) and a low value (e.g. 0) in order toproduce the output current I, that may also switch between a high andlow value following the switching of the control signal C.

In one or more embodiments, during an on-switch of the control signal C,the output current I may begin to rise from its low value, and, after arising transient, the output current I may remain at its high value.

The control signal C may remain at its high value for a time periodTpixel.

In one or more embodiments, at the end of the rising transient of theoutput current I, a time interval (e.g. Tpixel−T_(BSon)) may elapseduring which the output current I and the control signal C may remainhigh, while the first and second monostable signals BC1 and BC2 mayremain at the low values thereof.

In one or more embodiments, during an off-switch of the control signalC, the output current I may begin to fall from its high value, and,after a falling transient, the output current I may remain at its lowvalue.

The first BC1 and second BC2 monostable signals may be synchronized tothe control signal C: for example, also the monostable signals BC1, BC2may be synchronized to the switching of the control signal C. Forexample, to avoid requiring separate high frequency clocks for themono-stable, a delay chain may be used in order to provide such a highfrequency (in respect of Fpixel=1\Tpixel) controlling signals for theactive bootstrap.

In one or more embodiments, the first monostable (digital) signal BC1may switch to its high value when a rising edge of the control signal Coccurs, and may switch back to its low value after a first time intervalT_(BSon) has elapsed, that may substantially correspond to the risingtransient of the output current I; conversely, the second monostable(digital) signal BC2 may switch to its high value when a falling edge ofthe control signal C occurs, and may switch back to its low value aftera second time interval T_(BSoff) has elapsed, that may substantiallycorrespond to the falling transient of the output current I.

In one or more embodiments, the first time interval T_(BSon) and thesecond time interval TBSoff may be of equal duration.

Therefore, the active bootstrap switches, M4 or M5, may become active,i.e. conductive, when a high value of the respective mono-stablesignals, BC1 or BC2, occur, and remain non-conductive otherwise.

Accordingly, in one or more embodiments, the first active bootstrapswitch M4 may selectively connect the second intermediate node Y to thethird bias voltage node VB3, e.g. the second intermediate node Y and thethird bias voltage node VB3 may be short circuited when the first activebootstrap switch M4 is made conductive. As already discussed, the thirdbias voltage at the third bias voltage node VB3 may be provided as areplica of the voltage at the second intermediate node Y when thecurrent source transistor M1 is active, i.e. conductive.

Also, the second active bootstrap switch M5 may selectively connect thesecond intermediate node Y to the second bias voltage node VB2, e.g. thesecond intermediate node Y and the second bias voltage node VB2 may beshort circuited when the second active bootstrap switch M5 is madeconductive.

In one or more embodiments, at the beginning of an on switch, i.e. boththe control signal C and first monostable signal BC1 may be high, thevoltage at the nodes X and Y may be substantially the same (i.e. theelectronic switch M2 is conductive, short circuited), and the straycapacitors Cgd and Cgs may be discharged and charged, respectively, bythe current generated by the current source transistor M1.

Due to the stray capacitances, a high settling time may be needed forthe first and second intermediate nodes X and Y, and a high rising timewould result for the output current Iout. However, thanks to thepresence of the first active bootstrap transistor M4 being conductive(i.e. having current path being a short circuit), the first and secondintermediate nodes X and Y may be short circuited to the desired thirdbias voltage via the third bias voltage node VB3, which, as alreadydiscussed, may provide a low impedance replica of the DC voltage thatmay be present at the second intermediate node Y when the currentgenerator is active, i.e. conductive.

The third bias voltage may lead to a discharge and charge, respectively,of stray capacitors Cgd and Cgs at the beginning of the transient of thecontrol signal C.

Accordingly, the first monostable signal BC1 remains high (e.g. 1) forthe time interval T_(BSON) that may include one or more of:

-   -   the rising time of the control signal, and/or    -   a discharge time for the second stray capacitance Cgd, and/or    -   a charge time for the first stray capacitance Cgs.

In one or more embodiments, at the beginning of an off switch, i.e. whenthe control signal C falls to a low value and the second monostablesignal BC2 becomes high, the second active bootstrap transistor M5becomes conductive (i.e. the current path becomes a short circuit) andshort circuits the second intermediate node Y to the control terminal ofthe cascode transistor M3, thereby connecting the second intermediatenode Y to the second bias voltage node VB2 (e.g. short circuiting thefirst stray capacitance Cgs). Accordingly, the second active bootstraptransistor M5 may discharge the first stray capacitance Cgs at thebeginning of the falling transient of the output current I_(out),switching off the cascode transistor M3, and therefore the first straycapacitance Cgs is not discharged through the output current Iout(leading to a high settling time for the second intermediate node Y anda high falling time for the output current Iout).

Accordingly, the second monostable signal BC2 remains high (e.g. 1) forthe time interval T_(BSOFF) that may include one or more of:

-   -   the falling time of the control signal, and/or    -   a discharge time for the first stray capacitance Cgs, and/or    -   a partial or complete discharge time for the second intermediate        node Y.

In one or more embodiments, the source current transistor M1, theelectronic switch M2, the cascode transistor M3 and the two activebootstrap transistors M4, M5 may be field effect transistors, e.g.n-channel MOSFET transistors. Accordingly, the control terminal maycomprise a gate terminal and the current path plus first and secondterminals may comprise a source-drain channel plus source and drainterminals of the MOSFET. Also, in one or more embodiments, the bulk ofeach MOSFET transistor may be connected to ground GND.

FIG. 3 represents a non-limiting example of a device 100, which maycomprise one or more circuits 10 according to one or more embodiments.

For instance, according to a general layout which (with the exception ofthose elements which are discussed in detail herein) can be regarded asconventional in the art, the device 100 may comprise:

-   -   output driver channels comprising a circuit 10 according to one        or more embodiments, which may generate output currents applied        at output nodes IoutR, IoutG, IoutB and IoutR;    -   a digital interface 12, controlling the output driver channels        10,    -   a serial peripheral interface SPI 14,    -   an analog to digital converter ADC 16,    -   a temperature sensor 18,    -   a headroom detector 20,    -   a bandgap and references block 22,    -   a POR (power on reset) and supply monitor 24.

For example, the output currents at the output nodes IoutR, IoutG, IoutBand IoutR may be used to drive one or more pico-projectors D.

In laser diode driver application, in order to facilitate reducing powerconsumption, it may be desirable for the circuit 10 to operate atminimum headroom voltage for the current generator, in order tofacilitate minimizing the operative voltage on the output current nodeIout. Accordingly, the headroom detector 20 may be configured to receiveand measure the operative voltages on the output current nodes IoutR,IoutG, IoutB and IoutR of the output driver channels, in order tofacilitate adjusting such operative voltage in a dedicated feedback thatmay comprise an external voltage supply for the laser diodes D.

However, operating at low headroom voltages may lead to one or moredisadvantages, as previously outlined, e.g. devices operating in weakinversion and/or high values of stray capacitances.

One or more embodiments may thus present one or more advantages withrespect to the prior art, e.g. improvement of static performance:

-   -   power consumption: there may be no idle consumption insofar as a        complete switch off may be achieved in the off condition,    -   the possibility exists to manage a wide range output current        full-scale, while maintaining the described advantages.

Also, one or more embodiments may improve dynamic performance withrespect to the prior art:

-   -   it may lead to a faster rise time due to the first bootstrap        switch,    -   it may lead to a faster fall time due to the second bootstrap        switch,    -   it may lead to a faster settling time due to the circuit and the        introduced capacitors.

Also, one or more embodiments may facilitate obtaining such performancefor various (possibly all) values of the output current I_(OUT). Forinstance, according to one or more embodiments, the first and secondcapacitors C1 and C2 may have different values in order to be able toset a desired value depending on the output current Iout full scale.

One or more embodiments may thus relate to a circuit (e.g. 10) which maycomprise:

-   -   a first transistor (e.g. M1) and a second transistor (e.g. M3)        having respective control terminals and current paths        therethrough, the control terminal of the first transistor        coupled to a first bias voltage node (e.g. VB1) and the control        terminal of the second transistor coupled to a second bias        voltage node (e.g. VB2),    -   a first electronic switch (e.g. M2) having a selectively        activatable current path therethrough,

wherein the current paths through the first and second transistor andthe current path through the first electronic switch may be cascaded inan output current line between a circuit output node (e.g. Iout) andground (e.g. GND), the output current line having an intermediateportion (e.g. a portion of the output current line between the first andsecond intermediate nodes X, Y) between the first and second transistorwith a charge transfer node (e.g. Y) in the intermediate portion.

The circuit may comprise:

-   -   a second electronic switch (e.g. M4) arranged between a third        bias voltage node (e.g. VB3) and the charge transfer node, and    -   a third electronic switch (e.g. M5) arranged between the charge        transfer node and the control terminal of the second transistor,

the second electronic switch and the third electronic switch selectively(e.g. via signals BC1, BC2) activatable to a conductive state to providecharge transfer current paths coupling the charge transfer node to thethird bias voltage input node or the second bias voltage input node,respectively.

In one or more embodiments, the first electronic switch may be arrangedin the portion of the output current line between the first transistorand the second transistor.

In one or more embodiments, the first electronic switch may be arrangedin a respective portion of the output current line between the circuitoutput node and the second transistor or between the first transistorand ground.

In one or more embodiments, the circuit may comprise at least one of:

-   -   a first capacitor (e.g. C1) between the control terminal of the        first transistor and a node sourcing an inverted replica        (e.g. C) of the control signal, and/or    -   a second capacitor (e.g. C2) between the control terminal of the        third transistor and a node sourcing the control signal.

In one or more embodiments, the first and second transistor and thefirst, second and third electronic switches may comprise field effecttransistors, such as MOSFET transistors, having gate control terminalsand source-drain current paths therethrough.

Also, in one or more embodiments, the first and second transistor andthe first, second and third electronic switches may comprise fieldeffect transistors having their bulk terminals coupled to ground.

One or more embodiments may also relate to a device (e.g. 100) which maycomprise:

-   -   at least one circuit (e.g. 10) according to one or more        embodiments, which may be configured to provide an output        current (e.g. I) via the output current line between the output        node and ground of the at least one circuit, and    -   at least one driver circuit (e.g. 12) coupled to the at least        one circuit to provide the control signal to selectively        activate the first switch as well as activation signals (e.g.        BC1, BC2) to selectively activate the second switch and the        third switch.

One or more embodiments may also relate to a system which may comprise:

-   -   the device according to one or more embodiment, and    -   at least one user component, optionally a laser diode (e.g. D),        coupled to the device to receive therefrom the at least one        output current.

One or more embodiments may further relate to a method of driving any ofthe circuit, the device or the system of one or more embodiments, themethod may comprise:

-   -   selectively activating the first electronic switch to the        conductive state synchronously with an on-off control signal        (e.g. C) having rising edges and falling edges, and    -   selectively activating the second electronic switch and the        third electronic switch to the conductive state synchronously        with the rising edges and falling edges of the on-off control        signal selectively activating the first electronic switch.

Also, the method may comprise:

-   -   activating the second electronic switch to the conductive state        at the rising edges of the on-off control signal selectively        activating the first electronic switch, and    -   activating the second electronic switch to the conductive state        at the falling edges of the on-off control signal selectively        activating the first electronic switch.

The method may further comprise maintaining the second electronic switchand the third electronic switch to a non-conductive state over aninterval between the rising edges and the falling edges of the on-offcontrol signal selectively activating the first electronic switch.

Without prejudice to the underlying principles, the details andembodiments may vary, even significantly, with respect to what has beendisclosed by way of example only, without departing from the extent ofprotection.

The extent of protection is defined by the annexed claims. The claimsare an integral portion of the disclosure of the invention as providedherein.

The invention claimed is:
 1. A method, comprising: actuating a current source for a first time period to cause current to flow through a current path; selectively actuating a first switch to couple an intermediate node of the current path to a first bias voltage for a second time period beginning at a start of the first time period; and selectively actuating a second switch to couple the intermediate node of the current path to a second bias voltage for a third time period beginning at an end of the first time period.
 2. The method of claim 1, wherein the current flowing through the current path drives operation of a pixel.
 3. The method of claim 2, wherein the pixel includes a laser diode actuated in response to the current flowing through the current path.
 4. The method of claim 1, wherein the first bias voltage is less than the second bias voltage.
 5. The method of claim 1: wherein the second time period corresponds to a rising transient of the current flowing through the current path; and wherein the third time period corresponds to a falling transient of the current flowing through the current path.
 6. The method of claim 1, further comprising capacitively coupling a control signal for actuating the current source to a node receiving the second bias voltage.
 7. The method of claim 1, further comprising capacitively coupling a control signal for actuating the current source to a control terminal of the current source.
 8. The method of claim 7, further comprising applying the control signal to a control terminal of a switching transistor coupled in series with the current source along said current path.
 9. The method of claim 1, wherein selectively actuating the first switch is performed synchronously with the start of the first time period.
 10. The method of claim 1, wherein selectively actuating the second switch is performed synchronously with the end of the first time period.
 11. The method of claim 1, further comprising maintaining the first and second switches in a deactuated state between an end of the second time period and a beginning of the third time period while the current source remains actuated.
 12. A circuit, comprising: a current source circuit having a current path and configured to generate a drive current, said current source circuit actuated during a first time period, said current path having an intermediate node; a first switching circuit selectively actuated for a second time period beginning at a start of the first time period to couple the intermediate node of the current path to a first bias voltage; and a second switching circuit selectively actuated for a third time period beginning at an end of the first time period to couple the intermediate node of the current path to a second bias voltage.
 13. The circuit of claim 12, wherein the current source circuit comprises: a first transistor and a second transistor coupled in series to provide said current path and having respective control wherein a control terminal of the first transistor is coupled to receive a third bias voltage node and wherein a control terminal of the second transistor is coupled to receive the second bias voltage; and a third switching circuit coupled in series with the first and second transistors along said current path, wherein the third switching circuit is actuated by a control signal during said first time period.
 14. The circuit of claim 13, further comprising a driver circuit configured to provide the control signal and further provide actuation signals for selectively actuating the first and second switching circuits.
 15. The circuit of claim 12, further comprising a user component coupled to receive said drive current.
 16. The circuit of claim 15, wherein the user component is a laser diode.
 17. The circuit of claim 12, further comprising a capacitor coupled between a bias node of the current source circuit and an input configured to receive a control signal for actuating the current source circuit.
 18. The circuit of claim 12, wherein selectively actuating the first switching circuit is performed synchronously with the start of the first time period.
 19. The circuit of claim 12, wherein selectively actuating the second switching circuit is performed synchronously with the end of the first time period.
 20. The circuit of claim 12, further comprising maintaining the first and second switching circuits in a deactuated state between an end of the second time period and a beginning of the third time period while the current source circuit remains actuated. 